Device using a detection circuit to determine whether an output current thereof is source-induced or load-induced, and method therefor

ABSTRACT

A device which uses a detection circuit to determine whether an output current thereof is source-induced or load-induced, and the method therefor. The device which performs some type of operation based upon the determination as to whether the output current thereof is source-induced or load-induced, and method therefor. The detection circuit determines whether polarities of the output current and an output voltage are the same, and determines the output current to be source-induced if the polarities are the same and load-induced if the polarities are opposite each other. Such a device may have many applications, including use in systems where distinctions between source and load-induced currents are employed in feedback systems to control the system voltage source, systems where the system voltage source is not controlled, but other sources are controlled to influence a summation of voltages and currents at sensing locations, and systems for measurement instrumentation.

This application is a divisional of application Ser. No. 09/727,280,filed Nov. 30, 2000, now U.S. Pat. No. 6,531,898.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a device which uses a detection circuitto determine whether an output current thereof is source-induced orload-induced, and the method therefor, and more particularly, to adevice which performs some type of operation based upon thedetermination as to whether the output current thereof is source-inducedor load-induced, and method therefor. Such a device may have manyapplications, including use in systems where distinctions between sourceand load-induced currents are employed in feedback systems to controlthe system voltage source, systems where the system voltage source isnot controlled, but other sources are controlled to influence asummation of voltages and currents at sensing locations, and systems formeasurement instrumentation.

2. Description of the Related Art

An electronically programmable output impedance circuit is described inU.S. Pat. No. 5,708,379, issued Jan. 13, 1998 to Yosinski. Theelectronically programmable output impedance circuit therein is employedin a very specific manner in an AC source/analyzer product family, buthas broader application.

The AC source/analyzer product family includes models which areDC-coupled and which also employ a novel feedback circuit, an outputimpedance circuit, which is described in U.S. Pat. No. 5,708,379 issuedto Yosinski, which causes the source part of the product to exhibit acontrolled non-zero output impedance. The output impedance may be set tobe resistive or inductive. It may also be set to a complex value that isequivalent to series-connective resistive and inductive components. Themagnitudes of the resistive and inductive components are programmable.For realizations in the AC source/analyzer products, the resistivecomponent may be set to values between zero and one ohm, and theinductive component to values between 20 uH and 1 mH. Different rangesare possible subject to constraints imposed by necessity of maintainingstability in feedback loops.

DC-coupled members of the AC source/analyzer product family also employanother feedback circuit, a DC offset elimination circuit used as a DCservo control loop that may be enabled to eliminate unwanted DC offsetvoltages at the product's output.

As will be shown below, the output impedance circuit and the DC offsetelimination circuit, when active simultaneously, interact in anundesired manner that compromises the functionality and performance ofthe output impedance circuit at low frequencies including DC.

Practical applications for DC-coupled laboratory grade AC sourcesrequire simultaneous operation of both the output impedance circuit andthe DC offset elimination circuit mentioned above. When AC sources areused to simulate AC power systems, it is desirable to have bothresistive and inductive source impedances, since real systems exhibit afinite source impedance which includes both components. The magnitudesof the impedance components vary widely in real systems, makingprogrammability highly desirable.

Aside from the effects of source impedance, actual AC power systemsappear as nearly ideal sources for loads of the size that may powered byall but the very largest laboratory grade AC sources. To the extent thatthe source acts ideally, it will be capable of supplying any amount ofcurrent at any frequency. For this reason, it is essential for theoutput impedance circuit to work properly at very low frequenciesincluding DC. Loads with varying current consumption, for example, mayexhibit “beat-frequency” effects that produce AC power system currentsat low frequencies and/or DC. Adjustable speed drives (ASDs) are acommonly encountered example of such loads. Another common example is ahalf-wave rectified load which draws DC current from an AC power system.

Aside from the desired property of correctly simulating effects ofsource impedance and DC load current, it is otherwise essential to haveas little DC voltage present at the output of laboratory grade sourcesas possible since equipment with line-connected power transformers mayexhibit very little tolerance for DC voltage. DC levels of just a fewmillivolts can cause power transformers in the supplied equipment tosaturate.

On the other hand, it is undesirable for the source to actually beAC-coupled using, for example, an output transformer sincepractically-sized output transformers exhibit properties that precludeproper simulation of many events with DC content that occur in AC powersystems. Examples include partial cycle dropouts, non-symmetricalvoltage waveforms, etc.

To assist in an understanding of the present invention, it is helpful tounderstand the operation of the output impedance and DC offsetelimination circuits. In the discussion that follows, operation of thecircuits is considered independently and then in combination. Highlysimplified circuits imported from a circuit simulator and scaled tonominal values will be used to develop an understanding of the essentialconcept of the present invention.

FIG. 1A shows a very basic voltage source 100 which includes aninverting power amplifier 102. A DC voltage source 104 is shown torepresent an accumulated effect of undesired offset voltage sourcesencountered in practical devices. For the sake of simplicity, theoverall gain to the output of the basic voltage source 100 is set to −1.A current sensing element, or shunt, identified as 106, is coupled to1000× differential gain block 108 (hereinafter referred to as“differential gain block”) to provide a voltage output proportional tooutput current. Actual implementations may use differential amplifiers,but otherwise, the function and value for the current sensing element106 and associated differential gain block 108 are as might beencountered in practice.

Feedback is provided through an operational amplifier 110 which isconfigured as a unity gain follower. The voltage at the output side ofthe current sensing element 106 is connected to the positive terminal ofthe operational amplifier 110. The feedback sensing point is selected tocause voltage drops across the current sensing element 106 to be insideof the feedback loop, so that the voltage at the right hand, or output,side of current sensing element 106 is regulated by the action of thefeedback. A resistor 114 is connected between the output terminal of theoperational amplifier 110 and the negative terminal of the invertingamplifier 102. A resistor 116 is connected between the DC voltage source104 and the negative terminal of the differential amplifier 102. Acircuit comprising power amplifier 102, operational amplifier 110, andassociated input and feedback resistors 114, 116, and current sensingelement 106 are thus configured to function as an ideal voltage source.As an example, current sensing element 106, and resistors 114 and 116have the values 0.001 ohm, 10 k ohm and 10 k ohm, respectively.

Typically, another differential amplifier would be used in a voltagefeedback signal path, but for purposes of this example, it is useful toinclude the operational amplifier (configured as a unity gain follower)110 as shown since it is functionally transparent except for its actionto prevent current flowing in the feedback resistor 114 from becomingpart of the total current sensed by the current sensing element 106.Thus, only the load current is shown in meter 122. One volt displayed inmeter 122 corresponds to one ampere of load current. All of the meters120, 122, 124 shown in FIG. 1A are DC sensing.

A one-ampere current sink 118 is connected to the output node of thevoltage source 100 in FIG. 1A. The resulting current flow develops avoltage across current sensing element 106 which is amplified by theassociated differential gain block 108 and displayed as 1 volt in meter122. As would be expected with an ideal voltage source, the load currentproduces a negligible voltage drop at the voltage source output as shownby meter 120.

FIG. 1B shows essentially the same circuit as shown in FIG. 1A, but withDC input voltage source 104 set to −1 volt to represent significantaccumulated DC offset error voltage. In this case, an output voltage of+1 volt is now observed at meter 120, consistent with the operation ofthe overall circuit as a unity gain inverter. Since the current sink 118representing the load remains set to one ampere, the change in outputvoltage produces no additional current flow.

FIG. 1C shows the same circuit as shown in FIG. 1B, but with the currentsink 118 replaced by a resistor 132. In this example, the resistor 132has a value of one ohm. Consequently, one ampere of load current flows,as a result of the +1V output produced by DC input voltage source 104representing accumulated DC offset error voltages.

FIG. 1D shows the same circuit as shown in FIG. 1A with an additionalresistor 134 placed between the current sink 118 and a feedback sensingpoint 136 at the right-hand side of the current sensing element 106. Inthis example, the resistor 134 has a value of one ohm. As in FIG. 1A,the DC input voltage source is set to 0.0V. Note that the output voltageshown by meter 120 is now −1 volt, reflecting the voltage drop acrossthe resistor 134 induced by the load current. Note also that FIGS. 1Athrough 1D all show a voltage at the output terminal of invertingamplifier 102 which is 1 mV greater than the output voltage or, in thecase of FIG. 1D, than the voltage at sensing point 136. The differencereflects the voltage drop developed in response to the flow of oneampere through the resistance of 0.001 ohms for current sensing element(shunt) 106 as given for purposes of example.

FIGS. 1A through 1D depict basic configurations of a source part of anAC source/analyzer product, and show how internal DC offset voltageerrors are transferred to the output where they may produce undesiredeffects in a connected load. In addition, these figures help illustratean important distinction between output currents that are“source-induced” as compared to output currents that are “load-induced.”

The current in FIG. 1C is source-induced in the sense that an outputvoltage is required to produce current flow in a connected load. Theload in this case is represented by the resistor 132, but may be moregenerally represented by any complex impedance. In contrast, the currentin FIGS. 1A, 1B and 1D is load-induced in the sense that the currentflow is independent of the output voltage. More generally, this currentmay flow at any frequency including DC and, to the degree that thesource/sink representing the load is ideal, the current flow isindependent of and unaffected by changes in the source output voltage.

If the convention is adopted that current flow from the source into theload is considered “positive,” it may be observed from FIG. 1C that theoutput voltage producing the source-induced current has the samepolarity as the current. On the other hand, if the source has a finiteoutput impedance as represented by the resistor 134 placed outside thefeedback loop in FIG. 1D, load-induced currents will cause an outputvoltage that is opposite in polarity from the current flow.

In FIG. 1B the current polarity and output voltage polarity are thesame, but the output voltage is produced in response to DC input voltagesource 104, not in response to load current. This may be seen byreferring to FIG. 1A where DC voltage source 104 is set to 0.0V whilecurrent sink 108 remains set to 1 ampere, but in this case no outputvoltage results.

The significance of the distinction in polarities observed forsource-induced and load-induced currents in the presence of outputimpedance will be described in detail with regard to the presentinvention.

FIG. 2 shows the basic voltage source 100 shown in FIG. 1B with asimplified representation of a DC offset elimination circuit 200 addedthereto. The DC voltage source 104 has a value of −1 volt in thisexample. The DC offset elimination circuit 200 includes an operationalamplifier 202 which is configured as a differential integrator havingfor practical purposes infinite gain at DC and unity gain at a frequencydefined by a time constant for the RC elements comprising resistor 204,resistor 212 and capacitors 206 and 208. The values of the two resistorsare normally set to be equal; the same being true also for thecapacitors 206 and 208. The resistors 204 and 206 in this example havevalues of 100 k ohm and the capacitors 206 and 208 have values of 1 uF.For these values, unity gain occurs at 1.59 Hz.

Operational amplifier 202 and associated elements 204, 212, 206, and 208together compromise the differential integrator. A resistor 212 isconnected to the positive input of voltage follower 110 and to theright-hand side of current sensing element 106 (the output of thevoltage source 100). With this connection for resistor 212, thedifferential integrator comprised of amplifier 202 and associatedelements is configured to sample the output voltage (and DC contentthereof) of voltage source 100.

A resistor 210 has an end connected to the output of the amplifier 202(as well as an end of the capacitor 206). The resistor 210 has a valueof 10 k ohm in this example. The other end of resistor 210 is connectedto the junction of resistors 116, 114 and the negative input terminal ofthe inverting power amplifier 102. This connection to the “summingjunction” of amplifier 102 provides means for the differentialintegrator to introduce corrective feedback to eliminate undesired DCvoltages from the output of voltage source 100. As noted above, theconditions in FIG. 2 are as for FIG. 1B, that is, with DC input voltagesource 104 simulating accumulated DC offset error voltages set to −1V.This condition should produce +1 volt at the output of voltage source100, however, the action of the DC offset elimination circuit 200 servesto introduce corrective feedback via resistor 210 which removes DCvoltage from the source's output that otherwise would be observed giventhe DC input source.

From a frequency response standpoint, the action of the DC offsetelimination circuit 200 behaves as a first order high-pass functionresulting in a diminishing effect at higher frequencies. The −3 dbcorner frequency is effected by both the ratio of the feedback resistor210 to input resistor 116 and by the unity gain frequency of thedifferential integrator. When the resistor ratio is unity (as evidencedby both the resistor 210 and the resistor 116 having values of 10 k ohmin this instance), the integrator unity gain frequency equals the −3 dbfrequency for the overall system. At AC power system frequencies, theeffect of the DC offset elimination circuit 200 is sufficiently smallthat the AC gain error for voltage source 100 is less than 0.02% for ACsignals introduced at the location of voltage source 104 (the nominalinput for voltage source 100).

For a product designer, the circuit shown in FIG. 2 imposes a designtradeoff that must be appropriately resolved for an intendedapplication. If the corner frequency for DC offset elimination circuit200 is set too high, unnecessarily large AC gain errors result at powersystem frequencies, thus compromising the source's function as an idealAC power source. On the other hand, if the corner frequency is set toolow, the DC offset elimination circuit 200 will fail to adequatelyremove undesired error voltages at near-DC frequencies. Generally, thecorner frequency will be set as high as possible without unacceptablyimpacting gain accuracy at AC power system frequencies (e.g., at 50, 60and 400 Hz).

This distinction in frequency between desired and undesiredsource-induced signals is critically relevant for the present inventionas will be explained later.

FIG. 3 shows the basic voltage source 100 of FIG. 1B with an outputimpedance circuit 250. The output impedance circuit 250 includes aresistor 252 connected between the negative input terminal of poweramplifier 102 and the output of the differential gain block 108. In thisexample, resistor 252 has a value of 10 k ohm. In this case, only afixed resistive feedback loop is shown. Circuit values are scaled tocause output impedance, in this case resistance, of one ohm. In verysimple terms, the circuit acts to generate a feedback signal to summingjunction 254 (formed at the negative input terminal of power amplifier102 by the junction of resistors 116, 114 and 252) that is proportionalto output current. For the values indicated, one ampere of outputcurrent produces a one volt change in the output voltage thuscorresponding to one ohm of resistive output impedance. The outputvoltage shown by meter 120 is entirely due to the flow of currentthrough the output impedance since the DC error input voltage source isset to 0.0V. As shown previously in FIG. 1D, the output voltage isopposite in polarity from the output current according to the previouslyestablished convention that current flow from the source to the loadrepresents positive current flow. Opposition in polarity occurs because,as also noted previously, the current flow is load induced in thisexample.

The operational amplifier 110 has infinite input impedance because it isan ideal component such that no current flows in that path.

The differential gain block 108 senses the voltage developed acrosscurrent sensing element 106 in response to current, and develops anoutput voltage which is proportional to current, in this case, one voltis equal to one ampere. The voltage at the output of differential gainblock 108, when directed as feedback through resistor 252 has the effectof creating an output impedance. This result occurs because the sign ofthe feedback is selected such that the overall feedback to summingjunction 254 of inverting power amplifier 102 (including that providedvia voltage follower 110 and resistor 114) reaches equilibrium when theoutput voltage has in fact dropped slightly in response to outputcurrent. The combined action of the two feedback paths (one via resistor114 and the other via resistor 252) effectively creates the samebehavior at the output of voltage source 100 as would be observed withactual source impedance.

FIG. 4A shows the basic voltage source 100 with both the DC offsetelimination circuit 200 shown in FIG. 2 and the output impedance circuit250 shown in FIG. 3 added thereto with a −1V DC input voltage aspreviously shown in FIG. 1B.

In FIG. 4A, the DC error input voltage 104 is set to −1V. The DC offsetelimination circuit 200 correctly removes the effect of the DC errorvoltage from the source's output, but in addition acts to incorrectlyremove the output voltage that should result from one ampere ofload-induced current flowing through the one ohm output impedancesynthesized by the action of feedback of a signal proportional to outputcurrent via resistor 252. The true nature of this undesired effect ismore obvious when viewed in the frequency domain as shown in FIG. 4B.Substitution of a voltage controlled swept-frequency one ampere ACcurrent sink 260 for the fixed one ampere DC current sink 118 (see FIG.4A) as given in FIG. 4C permits the frequency response of the outputimpedance to be observed. Note that the meters shown in FIG. 4C are nowAC responding.

FIG. 4B shows the voltage appearing at the output of voltage source 100in response to the swept one ampere load-induced current produced by thevoltage controlled AC current sink comprised of current sink 260 and thecontrolling voltage source 262. From FIG. 4B, it may be seen that theload-induced current approximately produces the expected one volt at thesource's output for frequencies above 10 Hz, but if frequencies fallbelow 10 Hz, the expected output voltage is increasingly removed by theaction of the DC offset elimination circuit 200. The effect isequivalent to reducing the output impedance as the frequency approachesDC. This result is contrary to the desired effect since a resistiveoutput impedance should remain constant with respect to frequency. The−3 db point in the response is at 1.59 Hz due to the combined effect ofthe integrator time constant and the feedback ratio as describedpreviously. The phase angle at frequencies well below the −3 db pointreflects the fact that the impedance at those frequencies appearsinductive rather than resistive as desired. This outcome would beexpected from a magnitude response that increases by 20 db per decadewith increasing frequency, but frequency dependency remains undesired.

One of the uses for the AC source product shown in FIG. 4A is to conducttests of a wide range of electronic equipment for compliance withinternational standards for certain types of low frequency emissionsthat come from products, and in particular, one of these compliancetests is for a type of emission called flicker. Flicker is the varyinglight intensity observed when incandescent lamps are subjected tovarying AC voltages induced by time-varying currents in AC power systemshaving source impedance.

Many people think of an AC power, or mains, system as an infinitesource, in other words, as a voltage source with no source impedance. Infact, any mains system has a fairly significant amount of impedance.Accordingly, when conducting flicker tests it is necessary for thesource to exhibit an output impedance similar to that encountered inactual mains systems. For any type of equipment connected to the ACpower source, practical embodiments of AC source products such as shownin FIG. 4A attempt to simulate the source impedance of the main systemwhen conducting flicker tests.

There is generally a standard or reference value which is specified inthe emissions standard. For European 50 Hz compliance test standardsthis value is 0.4+j0.25 ohm, that is, 0.4 ohms resistive and 796 uH ofinductance. Different, but similar, values will typically be specifiedfor AC mains systems in other regions of the world. Further, it may bedesired to set the source impedance to a reference output impedancevalue for tests other than flicker compliance tests. This may be desiredsince similar values will be encountered in real mains systems and thushaving these values present allows the AC source to better simulate theenvironment within which the equipment under test (EUT) actually willoperate in practice.

Thus, AC source/analyzer products which are practical embodiments of thecircuit shown in FIG. 4A seek to simulate source impedance with afeedback loop. However, as noted above, there is another loop which isdirected towards removing a DC voltage from the output because an idealmain system has no DC content. Of course, a truly ideal mains system hasno output impedance either, but as described above source impedanceeither is required for conducting certain compliance tests or may bedesired for purposes of more accurately simulating actual mains systemenvironments. The problem that occurs with embodiments such as given inFIG. 4A is that the DC offset elimination loop interferes with theoutput impedance loop as you approach DC, and in fact, it totallycorrupts the same when you get to DC. Accordingly, there is no sourceimpedance at DC, but such a situation is not desirable since it does notproperly simulate the actual situation in mains systems. A pureinductance approaches zero impedance as the frequency approaches DC, butresistance is constant with frequency. As shown in FIG. 4B, theinteraction between the impedance loop and the DC offset removal loop issuch that the entire output impedance including the resistive componentimproperly approaches zero impedance as the frequency approaches DC. Forthe circuit values given as an example in FIGS. 4B and 4C, significanterrors begin to be introduced about 10 Hz.

One presumption is that if all else fails, output impedance may beimplemented by actually putting a discrete inductor and resistor inseries with the output of a product; in other words, by using physicalcomponents. If one does that, there is no problem with whether thesource works properly at DC. There are other problems, however. First,the resistive component dissipates a significant amount of power leadingto inefficient operation and possible design tradeoffs. Secondly, sinceit is desired to have a range of values available for both the resistiveand inductive components, advantages of cost-effective programmabilityattendant to a loop implemented approach as given in FIG. 4A are lost.It is possible to construct a programmable discrete component basedseries impedance, however, such implementations will be large, costly,and most likely unable to effectively simulate low impedance. This lastproblem occurs because parasitic resistance of switching elements suchas relays will be significant and variable with time therefore difficultto characterize and control yet these same switching components arenecessary to implement programmability.

As noted above, it is desired that the resistive component of the outputimpedance remain constant with frequency including DC. Further, it isdesired also that the inductive component should be an impedance whichis controlled solely by the selected inductance value and frequency andwhich is not influenced in addition by interaction with the DC offsetremoval loop. The reason why this behavior matters is that if there isan equipment connected as a test load which has a varying current, thecurrent is effectively amplitude modulated. Amplitude modulationproduces energy at side band frequencies about the carrier which in thiscase is the line frequency. These side bands can go down to DC forcertain types of equipment, so for purposes of characterizingload-induced voltage drops across the AC system impedance it is quiterelevant as to whether or not you have the proper impedance at DC.

One possible way to solve such a deficiency is by turning off the loopthat removes the DC, but then DC voltages would be present which is alsoundesired.

The above-described power source circuit is an AC source which is DCcapable. It is a switch mode inverter which provides an AC output atrelatively high power and simulates the AC power, or mains, system as asource, but is DC coupled.

The AC mains system is inherently AC coupled. There are transformers allthroughout the system and at multiple levels between low power loadsthat are plugged into a conventional wall outlet and the systemgenerator or source. Despite the fact that the system is not an idealsource for most low power equipment, the system looks like it is a verylarge source compared to the power that the load is drawing. Because thesystem is large compared to low power loads, it can support DC currentdrain from products without an apparent generation of DC voltage.Connected loads drawing DC current induce DC voltages in the sourcebecause of the source impedance, but this situation is quite differentfrom subjecting the load to a source-induced DC voltage. The reason forthis primarily has to do with transformers and the products themselves.If the product itself includes a power transformer and it's a relativelyhigh powered product, on the order of a kilowatt or so, a few millivoltsor a few tens of millivolts of DC can cause the product's powertransformer to saturate. Saturation produces abnormal operation in theload and for this reason it is very undesirable to have uncontrolled DCpresent in the source. Smaller products may be less susceptible tosaturation in the presence of a particular level of DC source voltage,but the risk of transformer saturation remains.

One of the problems with the situation described is that there is no wayto know what the load is. It may be large or small and it may or may notinclude a power transformer. The AC power source product is intended tobe general purpose and therefore capable of acting as a power source forvirtually any type of load. Thus, there is a situation where some loadswill be susceptible to the problems with DC content, and some won't, butthere is no way of knowing, which basically creates a situation where itis necessary to minimize source induced DC content. The problem is thatthe most effective way to do this because of practical constraints is toput a DC offset voltage elimination loop in the system. Considerationwas made of improving the quality of the components and the productsthemselves so as to eliminate the need for the DC offset eliminationcircuit loop or not use it when tests are being conducted where theoutput impedance presence is desired. However, such a modification wouldadd a significant amount of cost and not improve the results to asatisfactory level. Certainly, the benefit would not be to the degreethat would be achieved by keeping the DC offset removal loopoperational.

If both the DC offset elimination circuit and the output impedancecircuit are operating at the same time, the DC offset eliminationcircuit is incapable of distinguishing between low frequency AC and/orDC voltages appearing at the source output that are caused by the sourceas opposed to ones that are caused by the load. It is acceptable to havevoltages developed at the output that are caused by load behavior, butone does not want DC voltages at the output caused by source behavior.

SUMMARY OF THE INVENTION

Accordingly, it is an object of the present invention to overcome thedisadvantages of the conventional art.

It is another object of the present invention to provide a device whichdetects whether an output current is source or load-induced and toperform an operation based upon the detection.

It is another object of the present invention to provide a device whichdetermines an output current and an output voltage, and based upon thepolarities of the output current and the output voltage, determineswhether the output current is source or load-induced.

It is another object of the present invention to provide a voltagesource which detects and removes source-induced low frequency and DCoutput voltages and load currents thereby induced and maintainsload-induced low frequency and DC output voltages developed as a resultof load-induced currents flowing through an output impedance.

Additional objects and advantages of the invention will be set forth inpart in the description which follows and, in part, will be obvious fromthe description, or may be learned by practice of the invention.

The above and other objects and advantages are achieved by providing asource having an impedance and connected to a load, the sourcecomprising: a detection circuit to determine whether a current flowthrough the impedance is load-induced or source-induced; and aprocessing circuit to perform an operation based upon whether thecurrent flow is load-induced or source-induced.

The above and other objects and advantages are further achieved byproviding a power source circuit comprising: a voltage source togenerate an output voltage; a DC offset elimination circuit, which is aDC servo control loop connected to the voltage source, to eliminateundesired DC offset voltages added to the output voltage; and an outputimpedance circuit, which is a feedback loop connected to the voltagesource, to generate an output impedance for the source and whichoperates simultaneously with the DC offset elimination circuits; and adetection circuit, connected between the DC offset elimination circuitand the output impedance circuit, which determines whether a currentflow through the output impedance is load-induced or source-induced, andwhich selectively eliminates undesired voltages and resultant currentflow which are source-induced.

The above and other objects and advantages are still further achieved byproviding a method of controlling operations of a source which isconnected to a load, the method comprising: determining whether acurrent flow through an impedance of the source is load-induced orsource-induced; and controlling one of the operations based upon thewhether the current flow is load-induced or source-induced.

The above and other objects and advantages are still yet furtherachieved by providing a method comprising: generating an output voltagecontrolled by a feedback loop; forming a first additional feedback loopwhich eliminates DC offset voltages of the output voltage; forming asecond additional feedback loop to generate an output impedance for thepower source circuit simultaneously with the eliminating of the DCoffset voltages; and determining whether a current flow through theoutput impedance is load-induced or source-induced, and selectivelyeliminating the current flow which is source-induced.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other objects and advantages of the invention will becomeapparent and more readily appreciated from the following description ofthe preferred embodiments, taken in conjunction with the accompanyingdrawings of which:

FIG. 1A is a schematic diagram of a conventional voltage source;

FIG. 1B is a schematic diagram of the conventional voltage source shownin FIG. 1A, except with a different DC input voltage source simulatingthe accumulated effect of undesired DC offset error voltages;

FIG. 1C is a schematic diagram of a second conventional voltage source;

FIG. 1D is a schematic diagram of a third conventional voltage source;

FIG. 2 is a schematic diagram of a fourth conventional voltage source,in which the voltage source shown in FIG. 1B has a DC offset eliminationcircuit;

FIG. 3 is a schematic diagram of a fifth conventional voltage source, inwhich the voltage source shown in FIG. 1B has an output impedancecircuit;

FIG. 4A is a schematic diagram of a sixth conventional voltage source,in which the voltage source shown in FIG. 1B has both the DC offsetelimination circuit shown in FIG. 2 and the output impedance circuitshown in FIG. 3;

FIG. 4B is a graph of output voltage versus frequency of the voltagesource shown in FIGS. 4A and 4C, said voltage developed across thesource output impedance in response to load-induced current;

FIG. 4C is a schematic diagram of the voltage source shown in FIG. 4A,with a fixed DC current sink replaced with a voltage controlledswept-frequency frequency AC current sink, said reconfiguration of theload permitting examination of the voltage developed across the sourceoutput impedance versus frequency;

FIG. 5 is a generalized block diagram of a circuit for utilizing voltageand current polarities to distinguish between source and load inducedcurrent flows through an impedance placed in series between a voltagesource and a connected load;

FIG. 6 shows a first case for the circuit shown in FIG. 5, where thecurrent flow is source induced;

FIG. 7 shows a second case for the circuit shown in FIG. 5, where thecurrent flow is load induced;

FIG. 8A is a power source circuit with a corrective detection circuitfor a resistive output impedance component according to an embodiment ofthe present invention;

FIG. 8B shows the frequency response for a load-induced current of oneampere for the power source circuit shown in FIG. 8A.

FIG. 9A is a power source circuit with a corrective detection circuitfor an inductive output impedance component according to an embodimentof the present invention

FIG. 9B shows the frequency response for a load-induced current of oneampere for the power source circuit shown in FIG. 9A.

FIG. 10 is a schematic diagram of a realization of the power sourcecircuit shown in FIGS. 8A and 9A, which includes complex impedancefeedback loops for both resistive and inductive output impedancecomponents together with correcting feedback paths for both impedancecomponents;

FIG. 11A is a particular embodiment of the power source circuit shown inFIG. 10; and

FIG. 11B shows the performance of the circuit of FIG. 11A with resistiveand inductive components set to one-half scale (i.e., 0.5 ohm and 0.5mH)

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Reference will now made in detail to the present preferred embodimentsof the present invention, examples of which are illustrated in theaccompanying drawings, wherein like reference numerals refer to the likeelements throughout. The embodiments are described below in order toexplain the present invention by referring to the figures.

FIG. 5 is a generalized block diagram of a circuit for utilizing voltageand current polarities to distinguish between source and load inducedcurrent flows through an impedance placed in series between a voltagesource and a connected load. FIGS. 8A through 11B show a very specificuse of this technique in a feedback system to eliminate undesiredinteractions between other signals included in the same feedback system.The feedback system described in FIGS. 8A through 11B comprise a voltageprogramming source, various control loops and a power amplifier thesecomponents together comprising a laboratory grade DC-coupled AC powersource.

In FIG. 5, a voltage source 10 is shown with a connected load 20. Animpedance 12 and a current sensing element 14 are placed in seriesbetween the voltage source 10 and the load 20. The current sensingelement 14 provides information about the magnitude and polarity ofcurrent flows between the source and the load. A voltage sensing element16 is connected to the source side of the load 20 to sense voltagesdeveloped at the load 20 in response to current flow. The voltagesensing element 16 also provides magnitude and polarity information. Theseries impedance 12 causes voltages at the load 20 to differ from thevoltage source value for any non-zero current flow. Output signals fromthe current and voltage sensing elements 14, 16 are connected as inputsto a signal processing block 18, so that the signal processing block maybe configured for a wide variety of purposes.

If a convention is adopted that so-called “conventional” current flow(i.e., not electron flow) from the voltage source 10 to the load 20 isconsidered “positive,” then source induced positive current flow willproduce positive voltages at the voltage sense point, while load inducedpositive current flow will produce negative voltages.

FIG. 6 shows a first case, where because current flow is source induced,it follows that the load 20 is an impedance Z_(load). By Ohm's law,E_(load)=I_(load)*Z_(load).

FIG. 7 shows a second case, where, because the current flow is loadinduced, the load I_(sink), by definition, is a current source. Further,in order for current to flow from the source to the load (positivecurrent flow using the adopted convention), the load current must benegative in sign. More conventionally, the load must be a current sink.In this case:

E_(load)=E_(source)−I_(load)*Z_(series).

Setting E_(source) to zero:

E_(load)=I_(load)*Z_(series);

or:

−E_(load)=I_(load)*Z_(series).

From the equations given above, it is seen that the sign relationshipsbetween voltages developed across the load and current flow between thesource and the load may be used to determine whether current flow issource or load induced. Since the circuits described are linear,superposition causes multiple source and load effects to operateindependently, the overall effect in a system being the linear summationof individual effects. Further, it can easily be shown that the samesign relationships between voltage and current hold when current flowsfrom the load into the source, i.e. when the current flow is negativeusing the adopted convention. This being the case, the signrelationships described hold for any source or load induced current andfor any combination of source and load induced currents.

Returning for a moment to FIG. 5, it was previously noted that theseries impedance placed between the source and the load causes thevoltage E_(load) to differ from the voltage E_(source). Without thisimpedance, E_(load)=E_(sources) and it becomes impossible to determinepolarity relationships between E_(load) and I_(load). Accordingly, theremust be a non-zero impedance between the source and load for the conceptdescribed here to be effective.

From a theoretical standpoint, zero impedance is possible. In practice,it is difficult to achieve, in this case to advantage. Most practicalsituations will permit deliberate insertion of impedance or exploitationof unavoidable impedance.

There are many possible applications in which the polarity relationshipsdescribed above may be usefully employed. Three general cases aredescribed here:

1. In systems where distinctions between source and load inducedcurrents are employed in feedback systems to control the system voltagesource. The specific application described in FIGS. 8A through 11A tocontrol undesired interactions between feedback signals used tosynthesize output impedance and those used to eliminate DC offsetvoltages in laboratory grade AC voltage sources is an example of such asystem. 2. In systems where the system voltage source is not controlled,but other sources are controlled to influence the summation of voltagesand currents at the sensing locations. An example of such a system wouldbe an active harmonic filter that selectively opposes currents generatedby local harmonic sources while not generating signals to opposecurrents caused by harmonic voltage sources located elsewhere in thelarger system. 3. In systems where measurement is the objective ratherthan control. An example would be measuring instrumentation intended toprovide information about whether currents flowing at the sensing pointare caused by connected loads or by system voltage sources.

FIG. 8A shows a simplified implementation of a power source circuit forutilizing voltage and current polarities to distinguish between sourceand load induced current, according to the first case mentionedimmediately above. The power source has a circuit which is the same asthat shown in FIG. 4A, but with the addition of a detection circuit 270which acts to couple the output of the current sensing differential gainblock 108 into the DC offset elimination circuit 200. The detectioncircuit 270 comprises a resistor 272 which has one end connected toresistor 212 which in turn is connected to the positive terminal ofoperational amplifier 202. Operational amplifier 202 together withelements 204, 212, 206, and 208 comprises the differential integratorwhich provides corrective feedback to power amplifier summing junction254 via resistor 210 thereby to effect elimination of undesired DCoffset error voltages at the output of voltage source 100. The other endof resistor 272 is connected to the output of the differential gainblock 108. Note that in contrast to the operation of the power sourcecircuit shown in FIG. 4A, an output voltage of −1 volt is observed inmeter 120 connected at the output of the power source circuit shown inFIG. 8A. An output voltage of −1 volt correctly reflects the effect of aone ampere load-induced current flowing through an output impedance ofone ohm.

Recalling the earlier discussion noting the distinction between thesource-induced and load-induced output voltage changes, it may be seenthat the output voltage caused by the load-induced flow of one ampere inFIG. 8A is opposite in polarity from the current flow. If resistor 272and the resistor 212 are of equal value, the desired output voltage of−1 volt and the output of +1 volt from the current sensing differentialgain block 108 produce equal and opposite input signals into thedifferential integrator comprised of operational amplifier 202 andassociated elements 204, 212, 206 and 208. In this example, the value ofthe resistor 272 is 100K ohm which is equal to that of resistor 212. Inthe absence of a net input signal, the DC offset elimination circuitcomprises operational amplifier 202 and associated elements does notdevelop an output signal and therefore does not act to introduce acorrective signal via resistor 210 into summing junction 274 ofinverting power amplifier 102. Thus DC output voltage induced by loadcurrent is not removed. However, the effects of internal DC offsetsources represented by the input DC voltage source 104 are removed sincethere is no increase in load current and thus no current signal fromdifferential gain block 108 to compensate the DC voltage produced at theoutput by the source. Further, if a DC current sink 118 as shown in FIG.4 is replaced by a load impedance for the circuit shown in FIG. 8A, theoutput voltage and resulting source-induced output current would be ofthe same polarity, which would also result in the removal of the outputvoltage by DC offset elimination circuit 200 as desired.

FIG. 8B shows the frequency response for the power source circuitdepicted in FIG. 8A. An AC current sink 294 is shown in FIG. 8A torepresent a swept frequency AC current of one ampere drawn through theoutput impedance synthesized by the action of feedback via the outputimpedance circuit 250. Due to the aforementioned effects of thedetection circuit embodied by resistor 272, the output impedance remainsat a constant value of one ohm at all frequencies including DC.

In the discussion of the frequency response of the DC offset eliminationcircuit 200 shown in FIG. 2, it was noted that the designer needed toconsider the distinctions in frequency between desired source-inducedsignals and undesired source-induced signals. This distinction iscritically important to proper application of the power source circuitshown in FIG. 8A because the described enhancement to the power sourcecircuit is capable only of distinguishing between source-induced andload-induced signals. The enhancement provided by the detection circuit270 cannot by itself distinguish between desired and undesiredsource-induced signals, in particular within the frequency band in whichthe DC offset elimination circuit 200 has significant gain. Because theenhancement provided by the detection circuit 270 works via the DCoffset elimination circuit 200, its effect is also constrained to thissame frequency range. For the application described, desiredsource-induced signals fall within the frequency range where the gain ofthe DC offset elimination circuit 200 is reduced to the point of havingno effect. In other words, within the frequency range in which the DCoffset elimination circuit 200 and the detection circuit 270 hassufficient gain to have effect, all source-induced signals are undesiredand all load-induced signals are desired. The ability of the powersource circuit to distinguish between source-induced and load-inducedsignals is necessary within this frequency range, but all source-inducedsignals may be removed indiscriminately. The inability of the powersource circuit to distinguish between desired and undesiredsource-induced signals is irrelevant since this functionality isunnecessary for the intended application.

AC power system frequencies fall outside of the frequency range whereeither the DC offset elimination circuit 200 or the detection circuit270 have sufficient gain to have effect. Nor is there any need for themto have effect since there are no undesired source-induced signals ofconsequence to be removed at these frequencies. Therefore, the inabilityto distinguish between desired and undesired source-induced signals isof no consequence in this frequency range either, and the overalloperation of the power source circuit is as needed for a desiredapplication.

For these same reasons, the described power source circuit may not beapplied generally in applications where source-induced DC signals aredesired. However, it will in all likelihood be possible to deviseparticular schemes by which desirable source-induced signals aredistinguished from undesired source-induced signals with compensatinginputs to the DC offset elimination loop provided to preserve desiredsignals.

Thus far, the discussion is focused on the resistive component of theloop implemented output impedance. The same concepts may be applied tothe inductive component, however. As shown in FIG. 9A, the outputimpedance circuit 250 is replaced by the output impedance circuit 280.In other words, the feedback resistor 252 is replaced by a feedbackcapacitor 282. The capacitor 282 acts to differentiate the outputcurrent. The feedback signal is therefore proportional to di/dt and actsto produce the effect of an output inductance as described in U.S. Pat.No. 5,708,379 issued to Yosinski. Correspondingly, the detection circuit270 is replaced by a detection circuit 290, so that the resistor 272 isequivalently replaced by a capacitor 292. In this example, the value ofthe capacitor 282 is 100 nF and the value of the capacitor 292 is 10 nF.

FIG. 9A shows the power source circuit of FIG. 4A modified to introducean output impedance of 1 mH.

FIG. 9B shows the frequency response for a load-induced current of oneampere. As in FIG. 8A, an AC current sink 294 is shown in FIG. 9A torepresent a swept frequency load induced current of one ampere. Asdesired, the output impedance exhibits an inductive responsecharacteristic rising at 20 db per decade with the correct magnitude ofone ohm of inductive reactance at 159.15 Hz.

FIG. 10 shows a practical realization of an AC power source circuitaccording to the present invention which includes complex impedancefeedback loops together with correcting feedback paths into a DC offsetelimination integrator. In this example, a voltage source 300 has a DCoffset voltage circuit 400, an output impedance circuit 450 and adetection circuit 470 connected thereto.

The voltage source 300 has an AC input voltage source 304 which is shownin this example to reflect the intended function for voltage source 300in contrast to previous examples where the input voltage source wasgiven as a DC source to show injection of undesired DC offset voltages.Undesired DC offset voltage sources remain present in the system, butare not shown. A resistor 316 is connected between the AC input voltagesource 304 and a negative terminal of inverting power amplifier 302,said node identified as summing junction 374. A current sensing element306 is connected to the output of the inverting power amplifier 302, andultimately is connected to a load 340 for the AC power source circuit.

A differential voltage sensing amplifier comprising an operationalamplifier 310 and associated resistors 312, 314, 316, and 318 acts as adifferential amplifier with a gain of 0.01×. One end of a resistor 312is connected to the positive terminal of operational amplifier 310 andthe other end connected to the right hand side of the current sensingelement 306, said node being the output of voltage source 300.Connection of resistor 312 to the output of voltage source 300constitutes the positive input signal to the differential voltagesensing amplifier formed by operational amplifier 310 and associatedelements. The resistor 314 is connected between the positive terminal ofthe operational amplifier 310 and ground. The resistor 316 is connectedbetween a negative terminal of the operational amplifier 310 and itsoutput, and a resistor 318 has one end connected to the negativeterminal of the operational amplifier 310 and the other end connected tothe positive terminal of the inverting amplifier 302, which node is alsothe circuit “common” or ground for the entire system. This connectionprovides the negative input to the differential voltage sensingamplifier formed by operational amplifier 310 and associated elements.The resistor 320 is connected between the output of the operationalamplifier 310 and the summing junction 374. In this example, resistors312, 314, 316, 318, and 320 have values of 1M, 10K, 10K, 1M, and 10Kohms respectively.

The function of the voltage sensing amplifier formed by operationalamplifier 310 and associated elements is to sample the output voltage ofvoltage source 300 and to provide said sample as a “negative” feedbacksignal to summing junction 374, such feedback providing means by whichthe output voltage of voltage source 300 may be controlled and regulatedaccording to well understood principles. The attenuation factor of 100(gain of 0.01× set by the ratio's of resistors 314 and 316 to resistors312 and 318 respectively) provided by the voltage sensing amplifiercauses voltage source 300 to exhibit a voltage gain of 100 at its outputrelative to AC input voltage source 304. Accordingly, as an example, a3V r.m.s. signal at AC input voltage source 304 will cause a 300V r.m.ssignal at the output of voltage source 300. The output voltage forvoltage source 300 is developed between the node which is the junctionof resistors 306, 312, and the load 340 and the node which is thecircuit “common” or ground. The circuit “common” is connected to theother side of load 340, the positive input terminal of inverting poweramplifier 302, the “low” or common side of voltage source 304 and toother nodes within the system requiring connection to the system groundreference.

A differential current sensing amplifier comprising an operationalamplifier 330 and associated resistors 332, 334, 336, and 338 acts as adifferential amplifier with a gain of 5.657×. A resistor 332 isconnected between the positive terminal of operational amplifier 330 andground, and a resistor 334 connected between the same positive terminaland the output terminal of the inverting power amplifier 302. Connectionof the resistor 334 to the output terminal of inverting power amplifier302 constitutes the positive input signal to the differential currentsensing amplifier formed by operational amplifier 330 and associatedelements. A resistor 336 is connected between the negative and outputterminals of operational amplifier 330. A resistor 338 is connectedbetween the negative terminal of the operational amplifier 330 and theoutput end of the current sensing element 306, this connection providingthe negative input to the differential current sensing amplifier formedby the operational amplifier 330 and associated elements. The output ofthe differential current sensing amplifier is connected to and providesa sample of the output current to the output impedance circuit 450 andto the compensation circuit 420 which in turn is connected to the DCoffset elimination circuit 400. In this example, the current sensingelement 306 (a shunt) and resistors 332, 334, 336, and 338 have valuesof 0.01, 5.657K, 1K, 5.657K and 1K ohms respectively.

The function of the current sensing amplifier formed by operationalamplifier 330 and associated elements is to sample the output currentand to provide the sample to the output impedance circuit 450, where itmay be used to synthesize by means of feedback an output impedance forvoltage source 300. The sample is also provided to detection circuit420, where it may be used to detect output current polarity with respectto output voltage polarity thereby correcting the undesired interactionsbetween operation of the output impedance and DC offset eliminationcircuits previously noted. For the values given as an example for theelements 306, 332, 334, 336, and 338, the current sensing amplifierprovides an output signal of approximately 3.54 volts r.m.s. for anoutput current flow of 62.5 amperes r.m.s. (5 volts peak for an outputcurrent of 88.4 amperes peak).

The DC offset voltage circuit 400 comprises an operational amplifier 402and associated resistors 404 and 412 and capacitors 406 and 408.Together, operational amplifier 402 and associated elements act as adifferential integrator. A capacitor 408 is connected between thepositive terminal of operational amplifier 402 and ground, a capacitor406 connected between the negative and output terminals of operationalamplifier 402, and a resistor 404 is connected between the negativeterminal of operational amplifier 402 and ground. Resistor 412 isconnected between the positive input terminal operational amplifier 402and the output terminal of operational amplifier 310, said connectionproviding the input signal to the differential integrator formed byoperational amplifier 402 and associated elements. With connection tothe voltage sensing amplifier formed by operational amplifier 310 andassociated elements, means is provided by which the differentialintegrator may sample the output voltage of voltage source 300. Inaddition, a resistor 410 is connected between the output terminal of theoperational amplifier 402 and the summing junction 374, said connectionproviding corrective feedback to eliminate undesired DC offset and lowfrequency AC signals from the output of voltage source 300 as describedpreviously. In this example, the values of the capacitors 406 and 408are each 1 uF, and the resistors 404, 412 and 410 have values of 100K,100K and 10K ohms, respectively.

The output impedance circuit 450 is connected between the summingjunction 374 and the output terminal of the current sensing amplifier330.

The detection 470 is connected between the output terminal of thecurrent sensing amplifier 330 and the positive terminal of thedifferential integrator formed by operational amplifier 402 andassociated elements.

FIG. 11A is a particular embodiment of the power source circuit shown inFIG. 10, and is merely one embodiment thereof. The power source circuitshown in FIG. 11A includes both resistive and inductive feedback loopsmentioned above with the correcting feedback paths into the DC offsetelimination integrator. Since programmability is desired, a gain varyingcircuit 500 is introduced to allow control of the magnitudes of bothimpedance components. In this example, the output impedance circuit 450comprises a capacitor 454 and a resistor 452. In this example, thecapacitor 454 has a value of 17.68 nF and the resistor 452 has a valueof 56.57 k ohm. The gain varying circuit 500 comprises multiplying D/Aconverter 502 to vary the gain for the resistive component and amultiplying D/A converter 504 to adjust the gain of the inductivecomponent. Although desired for a variety of reasons, programmability isnot necessary to overcome the drawbacks of prior art systems.

The detection circuit 470 comprises a resistor 472 and a capacitor 474.The value of resistor 472 is 565.7 k ohm and the value of capacitor 474is 1.768 nF.

The corrective feedback into the DC offset elimination integrator 402via resistor 472 and capacitor 474 is taken from the output of the gainvarying circuit 500, thus ensuring that the correction signals properlyvary in proportion to programmed changes in the output impedance.

The generalized load 340 given in FIG. 10 is shown comprising a voltagecontrolled current sink 342 and a controlling voltage source 344 in FIG.11A. Together, elements 342 and 344 act to implement a swept frequencyAC current sink of one ampere, the purpose of which is to permitexamination of the output impedance versus frequency of the voltagesource 300.

With regard to FIG. 11A, the component values expressed above followfrom an actual realization in an AC source/analyzer having an outputvoltage capability to 300V r.m.s., output current capability to 62.5Ar.m.s., with programmable output impedance of 0-1 ohm resistive and 20uH-1 mH inductive. FIG. 11B shows the performance of the circuit of FIG.11A with resistive and inductive components set to one-half scale (i.e.,0.5 ohm and 0.5 mH). As would be expected, quadrature summation of thecomplex impedance at 159.15 Hz, where 0.5 mH equals 0.5 ohm, is 0.707ohm in magnitude. This impedance is reflected in FIG. 11B as an outputvoltage of 0.707 volts developed across the loop-implemented outputimpedance in response to a swept-frequency load induced current of oneampere.

Because resistor 472 introduces a DC-coupled signal path from thecurrent sensing element 306 to the summing junction of the DC offsetelimination integrator 402, it is essential that the signal path exhibitminimal DC offset errors as these will introduce uncorrected DC errorsat the source output. Offset sources at the input to the DC offsetelimination integrator 402 will appear at the output multiplied by afactor of 100×. These errors could defeat the purpose of the DC offsetelimination circuit 400.

In particular, 2-quadrant multiplying D/A converters are utilized inthis embodiment. Analog multipliers which might otherwise be usedexhibit large DC offset errors while multiplying D/A converters exhibitrelatively low DC offset errors that are primarily due to the offsetperformance of the voltage output operational amplifier and the effectsof leakage currents in the D/A proper. Both sources can be controlled toresult in very low DC offset errors.

As can be seen in FIG. 11A, the only other DC offset error source ofconsequence in the critical signal path is the current sensing amplifier330. This amplifier may also be selected to have very low offset errors.In the realizations employed thus far in the AC source/analyzerproducts, a single wide band current sensing differential amplifier isused. This amplifier has larger DC offset errors than would be the caseusing a narrow band, chopper-stabilized amplifier. Conflicts between theneed for bandwidth and DC accuracy could be resolved by using achopper-stabilized amplifier in parallel with the wide band amplifier.The chopper-stabilized amplifier's output would be used exclusively forthe output impedance circuit 450 and DC offset elimination circuits 400where wide band performance is not required.

With offset error sources controlled as described, either with orwithout use of the chopper-stabilized current sensing amplifier, DCerror is introduced by addition of the signal path through resistor 472and is acceptable for all but the most critical applications. Furtherimprovements in the performance of components in the DC-coupled path areanticipated in the near future. Other DC offset error sources in thesystem are much less easily controlled in practical realizations, soinclusion of the DC offset elimination circuit remains a practicalnecessity. Thus, the benefit of the present invention is relevant.

The above implementation is programmable so that any resistive valuebetween zero and 1 ohm and any inductive value between basically zeroand 1 mH may be obtained. Another advantage in using the above-describedloop implementation is use of small signal circuitry exclusively. Sincediscrete power resistors and power inductors are not involved, thefunction can very easily be made programmable. Programmability is highlydesirable because there are different standard values specified forvarious regions around the world to reflect differences in regionalmains systems. The advantage to a programmable solution is that it maybe readily re-configured to conduct a test for a different region of theworld. Another advantage of using the feedback loop implementation iselimination of discrete power components. A feedback loop basedimplementation is dissipationless, whereas in a discrete implementation,there is high current flowing through power components with theresistive components dissipating quite a bit of power, typically on theorder of several hundred watts.

Thus, with the addition of the detection circuit 470, the differentialintegrator 402 of the DC offset elimination circuit 400 is able todistinguish between a source induced current and a load induced current.By virtue of making this distinction the circuit is enabled to removethe source induced current by removing the voltage that produces it,thus removing both the source induced voltage and current, but does notremove the voltage developed across the output impedance in response toload induced current.

It may be considered that the detection circuit 470 is actually anenhancement of the DC offset elimination circuit 400 rather than anenhancement of the output impedance circuit 450 because the outputimpedance circuit 450 is not modified. It is performance of the outputimpedance circuit, however, which is corrected by virtue of the additionof the components of the detection circuit 470 feeding into the DCoffset elimination circuit 400.

FIG. 8A shows the output impedance circuit 250 as just a resistivefeedback loop, and FIG. 9A shows the output impedance circuit 280 asjust an inductive feedback loop, separately. FIG. 11A shows that both ofthese in a practical realization, rather than being fixed, are madevariable by feeding back the voltage that is proportional to outputcurrent through a gain adjusting circuit 500. Looking at FIG. 11A, thevoltage at the output of the current sensing amplifier 330 isproportional to current and is fed through the two multiplying D/Aconverters 502 and 504, these acting as digitally programmable gainblocks. The two multiplying D/A's vary the amount of voltage which isfed back through capacitor 454 and resistor 452, and that has the effectof varying the output impedance.

The aforementioned described power source circuit can be made to workarbitrarily close to DC depending on whether there is a frequencydifference between desired signals and signals not desired. The systemas shown loses functionality at DC for the reason that the detectioncircuit cannot distinguish between an undesired and a desired sourceinduced DC output. As noted previously, however, it may be anticipatedthat additional signals could be injected into the detection circuit topermit distinctions to be made between desired and undesired sourceinduced signals.

In FIG. 11A, the small signal AC source 304, which is a 60 Hz AC source,is input to the amplifier 302. There is a sufficient difference infrequency between the 60 Hz input signal and the 1.59 Hz cornerfrequency of DC offset elimination circuit 400 that the gain of the DCoffset elimination circuit is low enough to have no practical effect inremoving the desired 60 Hz signal. In other words, the difference infrequency between desired and undesired source induced signals may beexpressed as a ratio between a desired signal frequency and the DCoffset elimination circuit corner frequency. The value for the ratio isselected according to the accuracy with which the desired signals are tobe developed at the source's output. With a ratio of 50, for example,desired signals will be developed at the source's output with anattenuation of approximately −0.02% relative to values that would beexpected without the effect of the DC offset elimination circuit, othercircuit elements assumed to be ideal. This value for attenuation followsfrom the first order response characteristic of the DC offsetelimination circuit according to well understood principles.

It is possible to go arbitrarily low in frequency so that for purposesof example in this disclosure the corner frequency is set to around 1 Hzand the generator lowest frequency would be about 50 Hz. It would bejust as easily possible to make the AC source voltage 5 Hz, but if thisis done the DC offset elimination circuit corner frequency would need tobe set to a tenth of a Hz or thereabout to maintain similar accuracy(−0.02%) in generating the desired output signal.

The output of current sensing differential amplifier 330 is a voltageproportional to output current. The output of voltage sensingdifferential amplifier 310 is a voltage proportional to output voltage.The resistor 320 provides a feedback path for the voltage sensing signaldeveloped at the output of operational amplifier 310, this feedbackserving to regulate the output of the power source circuit internally.The resistor 412 provides the input to the DC offset elimination circuit400 whereby DC content at the source output may be sensed and eliminatedby the action of DC offset elimination circuit 400.

Capacitor 454 and resistor 452 are driven by the signal developed at theoutput of amplifier 330, said signal being the voltage proportional tooutput current.

The DC offset elimination circuit 400 and the output impedance circuit450 function on the summing junction 374, which is the input to thenegative terminal of power amplifier 302. Any signal that is fed back tosumming junction 374 is added to all other signals fed back there into,and the overall function of the entire system is controlled by thatfeedback.

In the power source circuit of the present invention, what is desired isto distinguish between low frequency AC or DC voltages that appear atthe output which are caused by the source as opposed to ones that arecaused by the load.

Another feature that is needed is to have output impedance all the waydown in frequency to DC. In practical terms, if there is an outputimpedance and one tries to draw current out of the source at DC, avoltage is going to develop across the impedance which means that a DCvoltage is present. In this case, the voltage is caused by the load, notby the source, and such a result is desirable. But the DC offsetelimination circuit removes this voltage as well as removing DC voltagesthat were induced by the source (which is the undesired behavior).

By adding a detection circuit as disclosed, it is possible todistinguish between source induced voltages appearing at the outputeither at DC or at very low AC frequencies and load induced voltagesappearing at the output either at DC or at very low AC frequencies.

The power source circuit according to the present invention allows boththe DC offset removal circuit and the output impedance circuit tofunction simultaneously by distinguishing between source induced DC andlow frequency AC output voltages and load induced DC and low frequencyAC output voltages.

Without benefit of the disclosed invention, the DC offset eliminationcircuit and the output impedance circuit would interact at lowfrequencies in a way that compromises the performance of the outputimpedance circuit. This effect is not significant at high frequencies,but at low frequencies where the DC offset elimination circuit isactive.

The second type of system described in relation to FIG. 5 is in thefield of shunt filtering of harmonic currents in AC power systems. Usein industry and elsewhere for this functionality has to do with the factthat many electrical and electronic products draw harmonic currents froma main system in addition to currents at the line frequency. Theelectricity supply utility industry does not wish to supply harmoniccurrents to connected loads, especially large industrial loads sincesuch currents contribute to loss and efficiency and other ill effects.The electric utility industry therefore is seeking regulation that wouldconstrain harmonic current “emissions”. At the present time there isconsiderable controversy surrounding this issue with utilitiesattempting to sponsor regulation that would force product manufacturersto redesign their products. Similar efforts are underway to establishrequirements for so-called “high impact” residential loads such astelevision receivers and personal computers. Not surprisingly,manufacturers are vigorously resisting these initiatives.

A compromise solution is to add what is called active filter technology.Active filters typically operate by drawing power at fundamentalfrequencies and re-injecting this power as reactive power at harmonicfrequencies in a manner which tends to counteract and cancel voltagesdeveloped at harmonic frequencies in response to load induced harmoniccurrent drain. One problem for systems of this type is to distinguishbetween harmonic voltages developed in response to locally connectedloads and harmonic voltages transported from elsewhere in the system. Anactive filter of practical size may be selected to compensate andcorrect for local effects, but cannot practically be sized to correctfor effects throughout the entire system.

As an example, if active filter technology is used in residences, thesame may be placed at the service entrance to the residence. The filtercontrol circuit would need to distinguish between harmonic currentsgenerated by equipment inside the residence and harmonic currentsbrought into the residence because of voltage sources in the utilitysystem. The “opposite sign effect” discussed above is usable todistinguish between source-induced current and load-induced current inthis application. This “opposite sign effect” relates to the fact thatwhen the output voltage is source induced, the sign or polarity of thevoltage and the current are the same, but when the voltage is loadinduced, the polarities are opposite. The voltages and currents are outof phase with one another from an AC standpoint. From a DC standpoint,they're just opposite in sign.

Undesired harmonic voltages are locally developed because of loadinduced harmonic current flowing through an impedance that isunavoidably present in practical utility distribution systems. Since theopposite sign effect is usefully present when source impedance ispresent, the disclosed invention may be used to distinguish betweenlocally induced currents (i.e. load induced) and currents flowing as aresult of source voltages (i.e. source induced). The difference infrequency between desired fundamental frequency signals and undesiredsignals at harmonic frequencies meets the remaining condition necessaryfor the disclosed invention to be usefully employed in this and similarapplications.

The example of active harmonic filtering in AC power distributionsystems is but one example of a system where the system voltage sourceis not controlled, but other sources are controlled to influence thesummation of voltages and currents at the sensing locations.

The third type of system described in FIG. 5 relates to a device wheremeasurements are the objective rather than control. As noted above, anexample would be measuring instrumentation intended to provideinformation about whether currents flowing at the sensing point arecaused by connected loads or by system voltage sources.

It should be noted that the values pertaining to the specific elementsthroughout the embodiments of the present invention have been includedmerely as examples, and the present invention should not be limited tosuch values. One of ordinary skill in the art would readily recognizethat other sets of values would still provide the benefits of thepresent invention. Further, the elements which form the variouscircuits, such as the voltage source, the output impedance circuit, theDC offset voltage elimination circuit and the detection circuit, havebeen disclosed merely as embodiments of these circuits, and the presentinvention should not be limited to such elements. One of ordinary skillin the art would readily recognize that other configurations of thevarious circuits would still make the present invention operational andprovide the benefits thereof.

According to the present invention, a device uses the concept of thepolarity of the output current versus that of the output voltage todetermine whether the current flow is load induced or source induced inthe fact of an impedance that causes an output voltage in response tocurrents that are load induced.

With regard to the power source circuit, a detection circuitdistinguishes between a source induced output voltage and a load currentinduced output voltage and corrects the operation of the DC offsetelimination circuit that removes the DC and low frequency AC componentsof the source induced output voltage. The specific application of theconcept is based on the same versus different polarities of the outputcurrent and output voltage. The difference in polarity is an actualoutgrowth of what occurs when there is a source with an impedance andthe current and output voltage are of the same polarity when the currentflow into the load is source induced and they are opposite in polaritywhen current flow is load induced.

The present invention detects the polarity of the output current as wellas the polarity of the output voltage and determines whether thepolarities are the same or opposite making a distinction whether thecurrent flow is load induced or source induced based upon the samenessor oppositeness of the polarities. Thus, the present invention makes adistinction between whether the current flow is source induced or loadinduced, and then controls operations of specific functions.

In the power source circuit, we want the load induced output voltage,but we want to eliminate the certain source induced voltages and loadcurrents dependent thereon. However, as noted above, there areapplications in which one would want to remove the load induced outputvoltage and not remove the source induced voltage, such as in a harmonicfilter that selectively opposes currents generated by local harmonicsources while not generating signals to oppose currents caused byharmonic voltage sources located elsewhere in a larger system. Further,in measurement instrumentation, it may be desirable merely to determinewhether the output current is source or load-induced, and neither thesource induced voltage nor the load induced voltage are removed.

What is claimed is:
 1. A power source circuit comprising: a voltagesource to generate an output voltage; a DC offset elimination circuit,which is a DC servo control loop connected to the voltage source, toeliminate DC offset voltages of the output voltage; and an outputimpedance circuit, which is a feedback loop connected to the voltagesource, to synthesize an output impedance for the power source circuitand which operates simultaneously with the DC offset eliminationcircuits; and a detection circuit, connected between the DC offsetelimination circuit and the output impedance circuit, which determineswhether a current flow through the output impedance is load-induced orsource induced, and eliminates the current flow which is source-induced.2. The power source circuit as claimed in claim 1, wherein the detectioncircuit comprises a resistor connected between the DC offset eliminationcircuit and the output impedance circuit, and the output impedance has aresistive component.
 3. The power source circuit as claimed in claim 1,wherein the detection circuit comprises a capacitor connected betweenthe DC offset elimination circuit and the output impedance circuit, andthe output impedance has an effective inductive component.
 4. The powersource circuit as claimed in claim 1, wherein the detection circuitcomprises: a resistor; and a capacitor; wherein the output impedance haseffective resistive and effective inductive components.
 5. The powersource circuit as claimed in claim 1, further comprising: a gain varyingcircuit which adjusts magnitudes of effective resistive and effectiveinductive components of the output impedance circuit.
 6. The powersource circuit as claimed in claim 5, wherein the gain varying circuitcomprises: a first multiplying D/A which varies a gain for the effectiveresistive component of the output impedance circuit; and a secondmultiplying D/A which adjusts a gain of the inductive component of theoutput impedance circuit.
 7. The power source circuit as claimed inclaim 1, wherein: the voltage source comprises: an inverting amplifierwhich inverts and amplifies an input voltage, a first differential gainblock to provide a voltage proportional to the output voltage of thesource which the proportional voltage is used as a negative feedbacksignal to control the source output voltage, a current sensing elementto sense a current output from the inverting amplifier, and a seconddifferential gain block to provide a voltage proportional to the currentpassing through the current sensing element; wherein the detectioncircuit is connected to an output end of the second differential blockgain.
 8. The power source circuit as claimed in claim 7, wherein: the DCoffset elimination circuit comprises: a differential integrator fromwhich a correcting signal to eliminate undesired DC and low frequency ACsignals is fed back to a summing junction of the inverting amplifier;wherein the detection circuit is connected to the input of thedifferential integrator.
 9. The power source circuit as claimed in claim8, further comprising a resistor connected between the differentialintegrator and the summing junction of the inverting amplifier.
 10. Thepower source circuit as claimed in claim 7, wherein the output impedancecircuit comprises: at least one resistive element to synthesize aresistive component in the output impedance, and one capacitor tosynthesize an inductive component in the output impedance, and eachconnected at a first end to the summing junction of the invertingamplifier and at a second end to the detection circuit and the voltageoutput of the current sensing differential gain block.
 11. The powersource circuit as claimed in claim 8, wherein the output impedancecircuit comprises: a first resistive element and a first capacitor andconnected at a first end to the summing junction of the invertingamplifier and at a second end to the detection circuit and the voltageoutput of the current sensing differential gain block.
 12. The powersource circuit as claimed in claim 11, wherein the detection circuitcomprises: a second resistive element and a second capacitor, whereinfirst ends of the second resistive element and the second capacitorconnected to the differential integrator and second ends of the secondresistive element and the second capacitor are respectively connected tothe first resistive element and the first capacitor of the outputimpedance circuit.
 13. A method of controlling a power source circuit,the method comprising: generating an output voltage; forming a firstloop which eliminates DC offset voltages of the output voltage; forminga second loop to synthesize an output impedance for the power sourcecircuit simultaneously with the eliminating of the DC offset voltages bythe first loop; and determining whether a current flow through theoutput impedance is load-induced or source induced, and eliminating thecurrent flow which is source-induced.
 14. The method as claimed in claim13, wherein the determining of whether the current flow through theoutput impedance is load-induced or source-induced comprises:determining polarities of the current flow and an output voltage acrossthe impedance; and determining the current flow to be source-induced ifthe polarities are the same and load-induced if the polarities areopposite to each other.